Professional audio equipment often employs electronically-balanced output circuits intended to mimic the behavior of output transformers as closely as possible. Such circuits are designed to accept a single-ended input voltage and to produce a differential output voltage with a low differential output impedance. They are further designed to possess a substantially higher common-mode output impedance (common-mode output impedance being defined as the impedance from either leg of the differential output to the ground or reference potential). This allows the differential output voltage to "float" with the common-mode voltage of the load, thus allowing the circuit to properly drive both balanced and ground-referred loads. This behavior is similar to that of an output transformer, wherein the differential output impedance is determined by the source impedance driving the primary reflected to the secondary (output) winding, while the impedance from either leg of the secondary winding to ground is quite high, being determined primarily by the stray capacitance from the secondary winding to ground. A consequence of this arrangement is that the output currents exiting the two legs of the balanced output are substantially equal in magnitude and opposite in polarity regardless of the load configuration.
A widely used circuit described in 1980 (T. Hay, "Differential Technology in Recording Consoles and the Impact of Transformerless Circuitry on Grounding Technique." Presented at the 67th Convention of the Audio Engineering Society, Journal of Audio Engineering Society (Abstracts), vol. 28, p.924 (December 1980)) is shown in FIG. 1. It accepts a single-ended input voltage v.sub.in with respect to ground at terminal IN. It produces a differential output voltage (equal to twice the input voltage) between nodes OUT+ and OUT-. This circuit accomplishes the desired goals with respect to differential and common-mode output impedances. Under normal operation, the differential output impedance is substantially determined by the sum of output resistances R.sub.O1 and R.sub.O2, as negative feedback around the operational amplifiers OA1 and OA2 substantially reduces their internal closed-loop output impedances. R.sub.O1 and R.sub.O2 are typically between 10 and 100 ohms in order to keep the differential output impedance relatively low. The common-mode output impedance is quite high, and can be infinite if the ratios of the resistances labeled R and 2R in the schematic are precisely maintained. It should be noted that mismatches in these resistor ratios can either reduce the common-mode output impedance if the mismatches are in one direction, or can lead to instability if they are in the other direction. This requirement for precise resistor-ratio matching is a drawback to this circuit.
It should be clear that the common-mode behavior of the circuit of FIG. 1 is governed by both opamps, OA1 and OA2. When driving a single-ended load, as in FIG. 2, the combined common-mode feedback forces the output currents to be equal and opposite (assuming exact resistor ratios around the opamps). This behavior is one of the most desirable properties of such circuits. However, if an input signal is applied to terminal IN that causes the output signal at the ungrounded output (in this case, OUT+), to exceed the maximum permitted by the power supply voltage, both the differential and common-mode feedback loops are broken. As is expected, the differential output voltage waveform at the OUT+ output would be "clipped" at the opamp's maximum output voltage. Its output current will be the output voltage divided by the load resistance. What is not as obvious is that, while clipping is occurring, the output current of the grounded OUT- output will be quite high, typically limited only by any protective current limiting circuit in the opamp, or by the maximum opamp output voltage divided by the value of the 10-to-100 ohm output resistor. This current must flow through an indeterminate path through the ground structure of the load device to return to the output stage, which can lead to disturbances on the audio waveform that are more audible than simple clipping.
An alternative approach to a floating balanced output circuit was described in 1990 by Chris Strahm in U.S. Pat. No. 4,979,218. Strahm's circuit includes separate feedback loops for differential and common-mode output signals. The differential loop is configured to force the differential output voltage to substantially equal the input voltage multiplied by some desired gain, and the common-mode feedback loop is configured to force the two output terminal currents to be equal and opposite. This at least opens up the possibility of preventing the clipping behavior and the audio waveform disturbances described above. Also, as described in the Strahm patent, precise resistor ratios are not necessary to maintain stability of the circuit.
Although not mentioned in the Strahm patent, in order to prevent a grounded output of such a circuit from going into current limiting when the active output is driven into voltage clipping, the common-mode feedback loop must remain active even though the differential feedback loop is disabled. In fact, the integrated circuit device manufactured by the assignee (Audio Teknology Inc.) based on the Strahm patent is implemented in a way that does not preserve the functionality of the common-mode feedback loop when the differential feedback loop is broken due to voltage clipping into a grounded load. As shown in FIG. 3, a differential pair of transistors, Q.sub.1 and Q.sub.2, accept the input signal and the differential feedback signal. Transistor Q.sub.3 provides the tail current I.sub.tail for the differential pair. Q.sub.3 is controlled by the common-mode feedback signal. In this case, the common-mode feedback signal is derived by sensing the sum of the output currents from the device, as described in the Strahm patent. Thus, the common-mode output voltage is adjusted via feedback through I.sub.tail until the two device output currents sum to nearly zero, and, are thus nearly equal and opposite. When voltage clipping occurs at either amplifier output, one of Q.sub.1 or Q.sub.2 will saturate while the other will be cut off. If the circuit is driving a ground-referred load from the output amplifier that is driven by the cutoff transistor, then there is no way for Q.sub.3 to affect the output voltage and common-mode feedback is also disabled. Without common-mode feedback to maintain control over the output currents, the grounded output amplifier conducts as much current as permitted by other aspects of the amplifier design, such as protective current limiting.